The present invention relates to a thin film resonator, more particularly to a thin film resonator filter circuit.
Thin film resonators (hereinafter xe2x80x9cTFRxe2x80x9d) are typically used in high-frequency environments ranging from several hundred megahertz (MHz) to several gigahertz (GHz). FIG. 1 illustrates a side view of a conventional TFR component 100. In FIG. 1, TFR component 100 includes a piezoelectric material 110 interposed between two conductive electrode layers 105 and 115, with electrode layer 115 formed on a support structure 120. The support structure 120 may be a membrane, or may be a plurality of alternating reflecting layers on a solid semiconductor substrate which may be made of silicon or quartz, for example. The piezoelectric material is preferably one selected from the group comprising at least ZnO, CdS and AlN. Electrode layers 105 and 115 are formed from a conductive material, preferably of Al, but may be formed from other conductors as well.
These TFR components are often used in filters, more particularly in TFR filter circuits applicable to a myriad of communication technologies. For example, TFR filter circuits may be employed in cellular, wireless and fiber-optic communications, as well as in computer or computer-related information-exchange or information-sharing systems.
The desire to render these increasingly complicated communication systems portable and even hand-held place significant demands on filtering technology, particularly in the context of increasingly crowded radio frequency resources. TFR filters must meet strict performance requirements which include: (a) being extremely robust, (b) being readily mass-produced and (c) being able to sharply increase performance to size ratio achievable in a frequency range extending into the gigahertz region. However, in addition to meeting these requirements, there is a need for low passband insertion loss simultaneously coupled with demand for a relatively large stopband attenuation. Moreover, some of the typical applications noted above for these TFR filters require passband widths up to 4% of the center frequency (for example, for a center frequency of 2 GHz, the bandwidth required would be about 80 MHz. This is not easily achieved using common piezoelectrics such as AlN, especially in combination with a plurality of reflecting layers on a solidly mounted substrate.
A conventional electrical circuit model for these resonators is illustrated in FIG. 2A. The circuit model is a Butterworth-Van Dyke model (BVD), and is comprised of a series RLC line which represents the motional (acoustic) resonance of the TFR between an input terminal 10 and output terminal 20. The series RLC line is in parallel with a capacitor CO representing the parallel plate capacitance (static capacitance) of the electrodes (for example, electrodes 105 and 115 in FIG. 1)
Impedance analysis of the BVD model illustrated in FIG. 2A yields a set of two resonant frequencies, a zero resonant frequency (xe2x80x9czeroxe2x80x9d) followed by a pole resonant frequency (xe2x80x9cpolexe2x80x9d). The separation of the zero from the pole is dependent on a piezoelectric acoustic coupling coefficient known as K2. This coefficient is a measure of how much of the acoustic energy is coupled into electrical, and varies with the piezoelectric material used in the TFR. As will be explained later, attempting to emulate standard LC bandpass filter design techniques is difficult because each TFR in the filter has the two resonances (pole and zero), whereas there is only one resonance in standard LC bandpass filter branches for common filter responses such as a Chebychev response, for example. As will be discussed later, this xe2x80x9cextraxe2x80x9d resonance (either the pole or zero resonance, depending on the design of the TFR) is disadvantageous in that it somewhat interferes with the passband as wider passbands are attempted. However, if materials with a larger K2 are used, this extra resonance will be farther away from the passband, which ultimately yields a wider bandwidth for the passband filter (the size of K2 is directly related to the amount of separation between pole and zero resonances).
A standard approach to designing filters out of resonators is to arrange them in a ladder configuration in a series-shunt relationship (i.e., a xe2x80x9cshuntxe2x80x9d resonator connected in shunt at a terminal of a xe2x80x9cseriesxe2x80x9d resonator). Each of the shunt and series resonators has a pole resonance and a zero resonance. To achieve a bandpass filter response, it is necessary to shift the pole frequency of the shunt resonator down in frequency to align somewhat with the zero frequency of the series resonator. This shifting of shunt resonator pole frequencies down in an attempt to match the series resonator pole frequency is typically accomplished by adding some material (such as a metal, metal oxide, etc.) to the top electrode of the shunt resonator.
Currently, the conventional way of designing TFR ladder filters is to design simple building blocks of TFR components which are then concatenated together (connected or linked up in a series or chain). In a simplified view, concatenation helps to achieve a larger stopband attenuation for the overall filter because each individual linked up section in the chain successively filters the signal more as it passes through the chain. FIGS. 2B and 2C illustrate the simple building blocks, which are commonly known as T-Cells and L-Cells. Referring specifically to FIG. 2B, a T-Cell building block 125 includes three TFR components 130A, 130B and 135. TFR""s 130A and 130B are xe2x80x9cseries armxe2x80x9d portions of the T-Cell block, being connected in series between an input port 132 and node 136, and node 136 to an output port 134 of T-Cell building block 125 respectively. Resonator element 135 comprises the xe2x80x9cshunt legxe2x80x9d portion of T-Cell building block 125, being connected in shunt between terminal 136 and ground. Similarly in FIG. 2C, an L-section block 145 used in a conventional TFR filter circuit includes TFR 146 comprising the series arm portion, with a TFR 147 connected in shunt to TFR 146 at terminal 144 to ground.
FIG. 3 illustrates a conventional TFR filter circuit. The filter circuit of FIG. 3 is created by concatenation of four T-Cells 151-154. As discussed above, the chaining up of a plurality of T-Cells provides a filter with high stopband attenuation. Further, redundant series resonators may be combined in order to reduce the size of the filter, as illustrated in FIG. 5. In FIG. 5, redundant components are shown combined (as compared to FIG. 3, series arm TFR components 160 and 165 are effectively xe2x80x9cpulledxe2x80x9d (i.e., combined) together to form one TFR series component 175) in an effort to save space. Although the xe2x80x9cinnerxe2x80x9d series branch electrode capacitances (see COS2 of TFR components 175, 180 and 185) in FIG. 5 are now different than the outer electrode capacitances (see COS of TFR components 190 and 195) because of the post-design combining, the xe2x80x9crootxe2x80x9d design is based on all of the series branch electrode capacitances being equal, and based on all the shunt branch electrode capacitances being equal, as illustrated in FIG. 3.
However, designing the filters illustrated in FIGS. 3 and 5 by the conventional concatenating approach has certain disadvantages. Namely, the filter designed by the above approach suffers poor flexibility in widening the passband width, as well as poor flatness performance when deviating from the center frequency. FIGS. 4A and 4B illustrate these effects in terms of insertion loss and return loss, as dB (y-axis) versus unit frequency (0.02 GHz/division, x-axis). FIG. 4A illustrates some common flatness and asymmetry problems. FIG. 4B illustrates common non-equiripple return loss.
FIGS. 6A and 6B illustrate the effects of using the conventional concatenation approach to widen bandwidth, depicting these effects in terms of insertion loss and return loss, as dB versus unit frequency. In order to attempt widening the bandwidth with the conventional concatenating approach, the shunt 5 resonator frequency set (pole and zero) must be shifted further apart from the series resonator frequency set. However, as the pole and zero resonant frequencies of the shunt resonator are shifted farther away from the series resonator frequencies, a xe2x80x9cbumpxe2x80x9d quickly forms in the return loss response in the center of the band. This can be graphically shown in comparing FIGS. 4B and 6B at the center of the passband (1.96 GHz), where return loss changes from  greater than 30 dB to about 10 dB. This bump is an indicator of how well the filter is xe2x80x9cunmatchedxe2x80x9d to 50 ohms (the standard for each T-cell building block in the filter), and the change in dB of the bump illustrates the drop off in impedance match. Moreover, because of the bump in the return loss response, a corresponding xe2x80x9cdipxe2x80x9d is formed in the insertion loss 15 response (see FIG. 6A). The primary reason for this is because all of the series resonators in the filter are resonant at the same frequencies, and all the shunt resonators in the filter are resonant at the same frequency.
Therefore, the conventional concatenation approach results in poor flatness performance and limited bandwidth. Although the filter formed by the concatenation approach is perfectly matched to 50 ohms at the center of the passband, the filter match deviates from 50 ohms more and more in approaching the edges of the passband and beyond into the stopband. Ideally, 50 ohm T-cell building blocks are concatenated in a proper manner only when they are matched adequately to 50 ohms over the entire band of interest. However, the concatenation approach shown in FIGS. 3 and 5 violate this principle by loading neighboring building blocks improperly as they deviate from the center of the passband, accounting for a less than ideal overall filter flatness performance and a pinching of the passband width.
Typically, when designing a filter as a whole, rather than by concatenating building blocks, a specific worst case return loss match performance is chosen. What typically happens is that the return loss (match) is better within the passband as compared to near the passband edges, where it falls off rapidly upon entering the stopband region. Furthermore, the filter is terminated by 50 ohm resistors on either side of it (which are 50 ohms regardless of frequency). In the case of concatenating several building blocks with similar match performance as above, each T-Cell would be terminating its neighboring T-Cell with a marginal return loss (match) near the passband edges, instead of a perfect 50 ohm match like at the center of the passband in the original case. Therefore, this phenomena tends to diminish the overall passband width for a given match performance criteria.
Accordingly, there is a need for a TFR filter circuit which can achieve better flatness performance in the passband, while achieving greater bandwidths.
The present invention provides a TFR filter circuit which may yield flatter passband responses and wider bandwidths as compared to those responses achieved by conventional TFR filter circuits. The TFR filter circuit has a plurality of TFR components arranged in a series branch-shunt branch relationship between input and output ports of the filter. Each of the TFR components in the series and shunt branches of the filter has a set of resonant frequencies, as well as a parallel plate electrode capacitance.
In one aspect, the present invention provides for the resonant frequency set of at least one TFR component in a series branch to be different from resonant frequency sets of other series branch TFR components in the circuit, and/or for the resonant frequency set of at least one TFR component in a shunt branch to be different from resonant frequency sets of other shunt branch TFR components in the circuit.
In another aspect, the present invention provides for the electrode capacitance of at least one TFR component in a series branch to be different from electrodes capacitances of other series branch TFR components in the circuit, and/or for the electrode capacitance of at least one TFR component in a shunt branch to be different from electrodes capacitances of other shunt branch TFR components in the circuit. The advantages of varying resonant frequency sets and/or electrode capacitance may enable wider bandwidth and improved flatness performance in the passband.